Linearization of an amplifier

ABSTRACT

The present invention relates to a method of linearization an amplifier ( 4 ) and also to amplifier circuitry. The method comprises the steps of generating a linearization signal ( 10 ) that substantially corresponds to low-frequency components of the square of an input signal ( 7 ) to the amplifier ( 4 ), processing the linearization signal, combining ( 15 ) the linearization signal with the input signal to form a modified input signal, and amplifying the modified input signal by means of the amplifier ( 4 ).

FIELD OF THE INVENTION

The present invention relates to linearization of an amplifier and inparticular, but not exclusively, to linearization of an amplifier foramplifying radio frequency signals.

BACKGROUND OF THE INVENTION

A communication system comprises signalling points or nodes, such asuser terminals, different exchanges, routers, switches, links, stationsand so on and an appropriate communication media between the signallingpoints. Signalling points may also be situated within an element of thecommunication system, wherein the communication may occur with theelement. The communication media may comprise, for example, a wiredinterface, a radio interface or an optical interface. The communicationmay be carried by analogue or digital signals or a combination of these,such as digitally-modulated analogue signals.

Amplification is required in various communication applications. Forexample radio frequency signals transmitted between signalling points ina radio communication system may need to be amplified during some stageof the transmission and/or reception. The signalling points may be, forexample, a transmitting station and a receiving station or anintermediate node of the communication system. The amplification of thesignals is required since the amplitude of a signal tends to beattenuated during the transmission of the signal between the signallingpoints, thereby decreasing the quality of the transmission. Also, noiseis added to the signal from other sources as well as from thetransmitting and receiving and the possible intermediate apparatusitself. The communication system is thus provided with amplifying meansto compensate for the attenuation and increase the signal-to-noise ratioof the signal by amplification of the signal.

Amplifiers intended to cover a range of frequencies should providelinear performance across the designated frequency band. Any amplifierintroduces linear, or AM-PM (amplitude modulation-phase modulation)distortion, where amplitude variations in the input signal causeundesirable phase variations in the output signal, and intermodulation,or AM-AM distortion, causing mixing between the different frequencycomponents present. An example of so-called third-order and fifth-orderdistortion components appearing around a simple two-tone signal atfrequencies f1 and f2 is shown by FIG. 1. Although other distortioncomponents are generated, these tend to be produced at frequencies thatare significantly higher than the desired signals, allowing their easyremoval with filtering. As illustrated, the magnitudes of thethird-order distortion components are normally greater than thefifth-order components. Even though not illustrated in FIG. 1,seventh-order (or even higher odd-order) distortion components may alsoappear around the carrier signals. However, the third-, and to a lesserextent, the fifth-order distortion will dominate the nonlinearity of anamplifier in most cases.

The non-linearity of an amplifier is caused by the finite output powerlimitation and non-ideal transfer function of the amplifier. Therefore,it is often desirable to provide amplifiers with some kind oflinearizing circuitry to reduce the distortion that is introduced. Astraightforward solution to the linearity problem exploits the fact thatthe non-linearity increases with the output power level of theamplifier. Thus, if the input level is reduced, or “backed-off”, theamplifier is arranged to operate only within its more linear region.However, this approach is not considered to be desirable in mostapplications as it fails to utilise the full range of available outputvoltage-swing and is therefore not power-efficient.

There are a number of well-established linearization techniques, whichhave been proposed over the years. One of the most well-known prior arttechniques is referred to as feedforward (F/F) linearization. Theconfiguration and basic operation of typical F/F circuitry is shown inFIG. 1. In the shown F/F arrangement an input signal consisting of twoclosely spaced tones is first sampled by a directional coupler in alocation before the amplifier. The separated fraction of the clean,undistorted input signal is passed through an amplitude andphase-shifting network while the input signal is passed through the mainamplifier. After the amplification, a further sample is separated fromthe amplified signal, said further sample including also the distortedcomponents of the amplified signal. The two sampled signals are combinedin a hybrid combiner in exact antiphase and with equal amplitudes inorder to cancel the original two-tone signal, leaving an “error” signalthat consists of only those distortion components that were introducedby the amplifier. The error signal is then amplified by an erroramplifier and subsequently combined with the distorted output of themain amplifier, again in precise antiphase and with equal magnitude,removing the extraneous distortion components and leaving a cleanamplified signal.

Feedforward linearizers are difficult to realize because all thecomponents operate at radio frequency (RF), and the phase and amplitudetolerances of the two cancellation loops are very tight and susceptibleto changes in temperature and ageing. In order to combat theaforementioned problems, a relatively complex control mechanism has tobe added in order to maintain acceptable performance. The feedforwardlinearizers are also temperamental, complex and inefficient to operate.Due to the above reasons, feedforward linearizers are relativelyexpensive and may thus be unsuitable for some applications requiringlinearization.

Another prior art linearization technique is predistortion. Inpredistortion the input signal is deliberately distorted prior to beinginputted to the amplifier in a manner that is contrary to thatdistortion that the signal will experience in the amplifier itself,resulting in a “cleaner” signal. Both analogue and digital predistortionhave been suggested. The predistortion systems may be constructed so asto form an open loop predistorter or a closed loop (i.e. adaptive)predistorter. The latter has the advantage of being able to adjust fordevice variations with temperature and time.

Predistortion linearizers operate to predistort the high-frequencycarrier itself, and therefore suffer many of the drawbacks of F/Flinearizers. The analogue predistortion linearizers operate only over arelatively small power range. The current digital predistortionlinearizers employ a complex structure, being difficult and expensive torealize. Despite the complexity, even the digital predistortionlinearizers have shown only a limited distortion improvement.

SUMMARY OF THE INVENTION

It is an aim of the embodiments of the present invention to address oneor several of the above problems.

According to one aspect of the present invention, there is provided amethod of linearizing an amplifier, the amplifier being provided foramplifying a signal, the method comprising steps of: generating alinearizing signal that substantially corresponds to the low-frequencycomponents of the square of an input signal to the amplifier; processingthe linearizing signal; combining the linearizing signal with the inputsignal to form a modified input signal; and amplifying the modifiedinput signal by means of the amplifier.

According to a further aspect of the present invention providescircuitry comprising: an amplifier provided with an input for receivinga signal; processing means for generating a linearizing signal thatsubstantially corresponds to the low-frequency components of the squareof an input signal to the amplifier and for processing the linearizingsignal with the input signal; and combiner means for combining thelinearizing signal with the input signal to form a modified input signalto be applied to the input of the amplifier for amplification.

In more particular embodiments of the invention, the generation of thelinearizing signal comprises squaring the baseband waveform of the inputsignal. Processing of the linearizing signal may comprise applyingamplitude adjustment and phase-shift to the generated linearizingsignal. Intermodulation distortion may be reduced at the output of theamplifier by means of addition of the linearizing signal to the inputsignal. The baseband waveform of the input signal may be obtained beforethe input signal is modulated and/or upconverted. At least a part of thegeneration and/or processing of the linearizing signal may beaccomplished by means of digital signal processing. At least oneanalogue to digital or vice versa conversion of signals may beaccomplished during the operation. The signals to be processed in thesignal processor and the produced linearizing signal may be intermediatefrequency signals. The baseband waveform of the input signal may besampled. A copy of the baseband waveform of the input signal may beprocessed during the operation.

The signal processor may be provided with feedback information from theoutput of the amplifier. The processing of the linearizing signal maycomprise conditioning of the linearizing signal based on said feedbackinformation. Amplitude and phase correction may be accomplished byemploying convolution to the time-varying waveform of the linearizingsignal. The generation of the linearizing signal may be dynamicallycontrolled.

The input signal may consist of two or more modulated carrier signals.The operation may comprise squaring two or more baseband waveforms andusing information associated with the mutual relations between saidcarriers. The information concerning the mutual relations comprisescarrier spacings between said one or more carriers.

The embodiments of the invention may enable efficient use of theamplifying range of an amplifier while reducing the intermodulation thedistortion present in the signal outputted from the amplifier. Theembodiments may enable an amplifying circuit with linearizationfunctionality that is less complex than that of F/F linearizers. Theembodiments may enable use of linearizing circuitry operating atsubstantially lower frequencies than those used in F/F linearizers. Thelinearizing circuitry components may also be realized digitally.Therefore, once an embodiment has been designed, the embodiment may bereproduced accurately. The linearizing circuitry designed in accordancewith the embodiments of the invention may also be substantially immuneto temperature changes and the passage of time, as well as being smallin size and power consumption.

BRIEF DESCRIPTION OF DRAWINGS

For better understanding of the present invention, reference will now bemade by way of example to the accompanying drawings in which:

FIG. 1 is a block diagram of a prior art linearized amplifier circuitry;

FIG. 2 shows a communication system in which embodiments of the presentinvention may be used;

FIG. 3 is a flowchart illustrating the operation of one embodiment ofthe present invention;

FIG. 4 is a schematic representation of an embodiment of the presentinvention;

FIG. 5 is a block diagram of circuitry in accordance with a preferredembodiment of the present invention;

FIG. 6 is a block diagram of circuitry in accordance with a furtherembodiment of the present invention;

FIGS. 7 and 8 illustrate further embodiments of the invention;

FIGS. 9 to 11 show experimental results obtained for the embodiment ofFIG. 3;

FIG. 12 is a block diagram of amplifier circuitry used for simulatingthe embodiment of FIG. 5; and

FIGS. 13 and 14 show the results obtained by the FIG. 12 arrangement.

DESCRIPTION OF PREFERRED EMBODIMENTS OF THE INVENTION

Reference will be first made to FIG. 2 illustrating a system in whichthe embodiments of the invention may be employed. The exemplifyingsystem is a cellular mobile radio communication system allowing aplurality of mobile stations MS1, MS2, MS3 to communicate with a base(transceiver) station BTS in a common cell via respective channels CH1,CH2, CH3. Although not shown, the mobile stations may also move from onecell to another cell. The radio communication between a transmittingstation and a receiving station may be implemented in any appropriatemanner and may be based on any communication standard. Therefore theradio link as such will not be described in more detail herein. Examplesof cellular communication systems include, without being limited to,standards such as AMPS (American Mobile Phone System), DAMPS (DigitalAMPS), GSM (Global System for Mobile communications) or various GSMbased systems (such as GPRS: General Packet Radio Service), CDMA (CodeDivision Multiple Access) or the proposed WCDMA (Wideband CDMA), UMTS(Universal Mobile Telecommunications System), IS-95 or IMT 2000(International Mobile Telecommunications 2000).

Third-order intermodulation distortion (IMD3) in an amplifier can besignificantly reduced by modifying the input signal by the addition of acarefully-engineered linearizing signal, produced from informationobtained from the signal to be inputted to the amplifier by appropriatemeans such as a signal-processing circuit or a data processing device.Steps for an embodiment for the linearizing are illustrated by the flowchart of FIG. 3. In the method a linearizing signal that correspondssubstantially to the low-frequency components of the signal that wouldbe obtained if the input signal were squared is generated and processed,preferably in a signal processor. The produced linearizing signal ispreferably substantially identical to the low-frequency portion of thesquare of the input, but may not be formed by actually squaring theinput signal or a copy or sample of the input signal. The generatedlinearizing signal may be processed further by conditioning thecomponents in an appropriate manner, as will be explained below. Thegenerated linearizing signal is subsequently combined with the inputsignal to form a modified input signal for the amplifier, i.e. a signalcontaining the original signal and the conditioned signal correspondingto the low-frequency portion of the square of the input signal. Themodified input signal is then fed into the amplifier and amplified. Asthe injected low frequency components will interact with the originalinput signal and the amplifier to cancel the third-order intermodulationdistortion generated by the amplifier, the distortion can be reduced atthe amplifier output. The following will outline in more detail both thefunctionality of this technique as well as some possible embodiments forimplementing the processing of the input signal information forgenerating the linearizing signal.

FIG. 4 shows a schematic block diagram for amplifying multi-tonecontinuous wave (CW) signals 1 by an amplifier 4 while employing theprinciples discussed above. Block 5 represents an arbitrary waveformgenerator (AWG) for outputting the low-frequency linearizing signal 10.The four adjacent RF signals t1 to t4 have a spacing S1 to S3 betweeneach other, respectively. The tones in this embodiment are continuouswaves with a periodic waveform (e.g. test signals having a sine waveformand carrying no information). Thus the low frequency portion of thesquare of the input CW signals will also be a periodic waveform,consisting of 6 second-order difference frequency components at thefrequencies S1 to S6. Therefore, once the periodic linearizing signal 10has been determined for a particular combination of carrier spacings andpower levels, it may be continuously outputted from the AWG 5 withoutadjustment. In a simple implementation the inherent second-orderdistortion of the amplifier itself is used to produce the square of theinput signal. In this case it is enough to digitally sample one periodof the periodic low frequency portion of the second-order distortioncomponents at the output of the amplifier 4 before the sample isprocessed and injected back to the input in a carefully-controlledmanner. The periodic low-frequency linearizing signal can be conditionede.g. by employing convolution or Fourier-Transform techniques, as willbe explained later in this description with reference to FIGS. 6 and 7,respectively. An arrangement constructed in accordance with the aboveprinciples for multi-tone CW input signals has also been tested, and thetest results will be discussed later with reference to FIGS. 8 to 10.

The proposed linearizing technique will now be discussed with referenceto a single carrier which has been subjected to modulation prior toamplification. A preferred embodiment will thereafter be discussed inmore detail with reference to FIG. 5. Examples of the single carrierapplications include handsets and mobile stations. It is appreciatedthat the embodiment described below can be employed to improve theoperating efficiency of amplifiers in other applications as well.

The analysis begins by representing the nonlinear input/outputcharacteristics (“transfer function”) of an amplifier by a simplethird-order power series of the form often employed:y(t)=ax(t)+bx ² (t)+cx ³ (t),  (1)

-   -   where y(t) represents the time-varying output signal, x(t) is        the time-varying input signal and a, b and c are the first,        second and third-order coefficients of the transfer function        respectively.

For the following exemplifying processing, a digitally-modulated carriersignal (designated by 7 in the FIG. 5) is represented in the frequencydomain as a carrier supplemented by a baseband spectrum (designated by 1in FIG. 5):x(t)

X(jω)=B(jω){circle around (×)}½[δ(ω−ω₀)+δ(ω+ω₀)],  (2)

-   -   wherein B(jω)) represents the baseband, ω is the independent        variable of angular frequency, ω₀ is the carrier frequency that        the modulated RF signal is centred at, ½[δ(ω+ω₀)+δ(ω−ω₀)]        designates a single sinusoidal frequency component at frequency        ω₀, and “{circle around (×)}” designates convolution of the        terms.

The second-order term, x²(t), of the nonlinear transfer functionproduces the following new output signal components from the above inputsignal shown:

$\begin{matrix}{\left. {x^{2}(t)}\Rightarrow{{X\left( {j\;\omega} \right)} \otimes {X\left( {j\;\omega} \right)}} \right. = {{B\left( {j\;\omega} \right)} \otimes {B\left( {j\;\omega} \right)} \otimes {1/{4\left\lbrack {{\delta\left( {\omega - {2\omega_{o}}} \right)} + {\delta\left( {\omega + {2\omega_{o}}} \right)} + {2{\delta(\omega)}}} \right\rbrack}}}} & (3)\end{matrix}$

The third-order term of the transfer function, x³ (t), produces thefollowing new output signal components:

$\begin{matrix}{\left. {x^{3}(t)}\Rightarrow{{X\left( {j\;\omega} \right)} \otimes {X\left( {j\;\omega} \right)} \otimes {X\left( {j\;\omega} \right)}} \right. = {{{B\left( {j\;\omega} \right)} \otimes {B\left( {j\;\omega} \right)} \otimes {B\left( {j\;\omega} \right)} \otimes {1/{8\left\lbrack {{\delta\left( {\omega - {3\omega_{o}}} \right)} + {\delta\left( {\omega - {3\omega_{o}}} \right)} + {3{\delta\left( {\omega - \omega_{o}} \right)}} + {3{\delta\left( {\omega + \omega_{o}} \right)}}} \right\rbrack}}} = {{{B\left( {j\;\omega} \right)} \otimes {B\left( {j\;\omega} \right)} \otimes {B\left( {j\;\omega} \right)} \otimes {1/{8\left\lbrack {{\delta\left( {\omega - {3\omega_{o}}} \right)} + {\delta\left( {\omega + {3\omega_{o}}} \right)}} \right\rbrack}}} + {{B\left( {j\;\omega} \right)} \otimes {B\left( {j\;\omega} \right)} \otimes {B\left( {j\;\omega} \right)} \otimes {3/{8\left\lbrack {{\delta\left( {\omega - \omega_{o}} \right)} + {\delta\left( {\omega + \omega_{o}} \right)}} \right\rbrack}}}}}} & (4)\end{matrix}$

The third-order distortion appearing around the carrier signal isdesignated in the above by the underlined terms in the third-orderexpansion (4).

The linearising signal consisting of the low-frequency portion of thesquare of the input signal that is inputted to the amplifier along withthe carrier signal to reduce the third-order distortion can berepresented in the frequency domain as:I(jω)=αe^(−jθ) [B(jω){circle around (×)}B(jω){circle around(×)}δ(ω)],  (5)

-   -   wherein αe^(−jθ) bulk amplitude and phase-shift term determined        by the injection-signal processing means.

If the above “injection” signal is examined, it can be seen to consistof the frequency-domain convolution of the baseband spectrum withitself. The time-domain equivalent of convolution is multiplication, sothe injection signal can be formed simply by squaring the time-domainbaseband waveform and applying prescribed amplitude and phase-shift.Thus, in order to generate this linearising signal it is not necessaryto square the whole RF input signal, it is sufficient to square thebaseband signal alone. If the injection signal is added to the originalmodulated carrier that is centred at frequency ω₀ the input signal andthe components generated at the output of the amplifier are changedaccordingly:X′(jω)=X(jω)+I(jω)=B(jω){circle around (×)}½[δ(ω−ω₀)+δ(ω+ω₀)]+αe^(jθ)[B(jω){circle around (×)} B(jω){circle around (×)}δ(ω)]  (6)

-   -   Second-order distortion components:

$\begin{matrix}{{{X^{\prime}\left( {j\;\omega} \right)} \otimes {X^{\prime}\left( {j\;\omega} \right)}} = {2{{B\left( {j\;\omega} \right)} \otimes {B\left( {j\;\omega} \right)} \otimes {1/{4\left\lbrack {{\delta\left( {\omega - {2\omega_{o}}} \right)} + {\delta\left( {\omega + {2\omega_{o}}} \right)} + {2{\delta(\omega)}} + {\alpha\;{\mathbb{e}}^{{- j}\;\theta}{{B\left( {j\;\omega} \right)} \otimes B}{\left( {j\;\omega} \right) \otimes B}{\left( {j\;\omega} \right) \otimes \left\lbrack {{\delta\left( {\omega - \omega_{o}} \right)} + {\delta\left( {\omega + \omega_{o}} \right)}} \right\rbrack}}} \right.}}}}} & (7)\end{matrix}$

There are now new third-order distortion components produced by thesecond-order term of the nonlinear transfer-function, as underlined inthe above equation (7).

If the underlined second-order term in equation (7) is compared with theunderlined third-order distortion around the carrier in the third orderequation (4) it can be seen that the second-order amplifier nonlinearity(7) now produces new third-order distortion components around thecarrier, identical to those already being produced by the third-orderterm (4), except for an additional amplitude and phase-shift termαe^(−jθ). Hence, if the amplitude α is chosen so that the amplitudes ofthe two sets of underlined components in equations (4) and (7) areequal, and the phase-shift θ is chosen such that the phases of saidcomponents are opposite, distortion cancellation will occur.

Reference will now be made to FIG. 5, which shows a block diagram of apreferred embodiment of the invention. The embodiment may beimplemented, for example, in the base transceiver station BTS and/or themobile stations MS of FIG. 2. A dedicated digital signal processor (DSP)circuit 5 shown in FIG. 5 may be used to generate the requiredlinearizing signal from the baseband waveform that will subsequentlymodulate the RF carrier to form the input signal. The processing circuit5 will produce the linearizing signal 10 by squaring the basebandwaveform and applying bulk phase and amplitude adjustment to it. Thismay be operated under control of a control unit 21. The block diagramillustrates a baseband waveform 1 being modulated onto a carrier signalto be inputted into an amplifier 4. The amplifier may be any type ofamplifier, such as a radio frequency power amplifier (RF PA) or asmall-signal amplifier. The baseband signal may be any signal that maybe used for the transmission purposes such as a digital signal or ananalogue signal.

Before the baseband signal is inputted into the amplifier, somepre-processing or pre-conditioning of the signal may be performed. Forexample, the baseband signal (i.e. the information to be carried) mustbe translated upwards in frequency in order, among other reasons, toenable it to be transmitted through the air and to prevent interferencewith other transmissions. In FIG. 5 the upwards translation is performedwith an upconverter 12 which is arranged to convert intermediatefrequency (IF) signals into radio frequency (RF) signals in a knownmanner. The information signals may be any signals carrying information,such as waveforms carrying, for example, audio, visual or digitalinformation that is translated into electrical signals by transducerssuch as microphones or cameras or data processing devices.

In addition to upconversion, in order to make efficient use of frequencyspace, modulation is used in most applications to vary the amplitude,frequency and/or phase of an RF carrier. These pre-processing methodsare often referred to as amplitude modulation, frequency modulation andphase modulation, respectively. The signal path to the amplifier 4 maythus include means for processing the carrier signal prior to the signalbeing inputted into the amplifier. The FIG. 5 arrangement comprises amodulator 3 for the modulation of the carrier signal as discussed above.The input signal is also shown to be passed through a time delay means 2during the pre-processing of the signal, prior to the inputting of thebaseband signal into the modulator 3. The time delay means 2 may berequired because the conditioning that may be performed on the injectionsignal within the circuit 5 will take a finite amount of time. If thisprocessing time is such that the linearizing signal does not arrive atthe amplifier 4 correctly time synchronized with the input RF signal,the two sets of third-order distortion components already discussed willnot coincide in antiphase, and distortion reduction will not occur. Thetime delay may thus be required to compensate for this problem. Theposition of the time delay can be anywhere between the point where thebaseband signal is split (14) and the amplifier (4). Also, the timedelay 2 may not be required at all if the processing in circuit 5 can beperformed rapidly enough.

In addition to the exemplifying pre-processing shown by FIG. 5, thereare also other possibilities to process or condition the input signalprior to inputting the carrier signal into an amplifier, such aspre-amplification or filtering. However, as the possible pre-processingtechniques are known and do not form a part of the invention as such,they will not be discussed in more detail herein.

After the pre-processing means, the modulated and upconverted signal isinputted into the amplifier 4. The amplified signal is outputted andcarried further on a signal path 9 from the output of the amplifier 4 ina known manner.

The circuitry of FIG. 5 includes further a digital signal processing(DSP) circuit 5 in accordance with a preferred embodiment for generatinga linearizing signal for use in linearization of the amplifier 4. Theprocessing circuit 5 is arranged to process information it receives viaconnection 8 from the baseband signal 6 prior to the baseband signalbeing processed by the pre-processing means. The received informationcontains information of the baseband, such as the waveform of thebaseband, before the baseband is subjected to pre-processing such asupconversion. The baseband information may be obtained at a point 14located in the signal path 6.

It is to be appreciated that the information may be obtained elsewherethan at the point 14. For example, the IF output of the modulator 3 maybe split, and one half thereof may then be downconverted to produce thebaseband again, which may then be squared and processed further in thecircuit 5.

The processing or linearizing circuit 5 includes means 23 for providinga squaring functionality. For example, the squaring functionality may beimplemented in the DSP block 5 as a mathematical function (i.e. bymultiplication of a signal with itself), or, in an analogueimplementation, e.g. by means of a diode. Phase-shift means 25 andamplitude-adjustment means 24 are also provided for adjusting the bulkamplitude and phase of the digital signal. These may be located in theinjection-signal processing means at any position between the points 14and 15.

The signal processing circuit 5 may also be provided with avariable-gain amplifier 22 or other suitable means, which is arranged toadjust the amplitude of the received information signal. Anothervariable-gain amplifier 24 or similar means may be provided between thesquaring means 23 and the phase-shift means 25, or at any positionbetween the squaring means 23 and the combining means 15. In addition,the circuit 5 may comprise a digital filter 26 for filtering thegenerated signal prior to it being outputted from the circuit 5.

In the preferred embodiment the processing circuit 5 is a digital signalprocessing circuit. Thus, if the information from the baseband (e.g. thewaveform or logical state of the baseband) is provided in an analogueform and the processing circuit comprises digital signal processing, itmay be necessary to provide an analogue to digital converter (ADC) 20for converting the information signal into digital form. However, theinput information signal 8 may already be in a suitable format fordigital processing, whereby the converter 20 would not be required.Correspondingly, if the processing block 5 is implemented digitally adigital to analogue converter 27 or other suitable means may be requiredfor converting the generated linearizing signal into a format suitablefor combining with the carrier signal.

It is possible, though not necessarily essential, to provide filteringor other suitable means 26 (digital or analogue) to predistort theinjection signal in order to compensate for the effects of any filteringand/or coupling components that may be used in order to combine theseparate RF carrier 7 and intermediate frequency (IF) linearizing signal10.

Although not necessary in all implementations, the operation and variousfunctionalities of the processing circuit 5 may be controlled by thecontrol unit 21. The control unit 21 may be arranged to adjust thevarious components of the circuit 5 in order to maintain correctoperation. The control unit may use external information for the controlof the generation and conditioning processing. The other information mayinclude information such as carrier spacing in multicarrierapplications, transmission power levels, temperature and so on. An input28 to the controller represents this “external” information. The controlunit is not necessary in all embodiments, and may thus be omitted, inwhich case the linearizer may be referred to as being “open-loop”.According to one possibility the control unit 21 provides at least someof the functionalities of the linearising signal generation andprocessing components of the circuit 5. In addition, the control unit 21may be arranged to process possible information provided by a feedbackconnection 11. An example of the possible feedback control operationswill be discussed in more detail later in this description.

In FIG. 5 the linearizing signal 10 is combined with the radio frequencycarrier signal at a combination (or “injection”) point 15 before theradio frequency signal is inputted into the amplifier 4. In someembodiments it may be necessary to provide low-pass filtering means 13in the signal path between the linearizing signal generating circuit 5and the injection point 15 in order to prevent high frequency signals online 7 from entering the circuit 5 and to minimize the effect of theconnection 15 on the RF signal path 7.

According to one possibility the linearizing signal is injected to theamplifier via the DC power-source connection (typically an inductive“bias-feed line” or an inductor having a high impedance at the RFfrequency) at the input of the amplifier. In this embodiment the powersupply input provides a pre-existing low-pass filtering network.

The above described arrangement facilitates use of low frequencycomponents in the linearizing circuit 5, since the frequency of thebaseband information signals transmitted from the detection point 14 tothe linearizing circuitry signal generating and processing circuit 5 maybe of a considerably lower frequency than the RF carrier signals to beinputted into the amplifier 4 (e.g. at frequencies in the order of kHzor MHz).

Although the above discusses the third-order distortion, the embodimentis analogously applicable in linearization of the fifth-order or evenhigher order distortion with this technique. The experimental resultsthat will be discussed below have shown improvement in fifth-orderdistortion as well. The fifth-order distortion also appears around thecarrier, but is not as problematic as the third-order distortion in mostinstances, which tends to dominate.

Embodiments of the invention may also be used to linearize amulti-carrier power amplifier. An example of the possible multi-carrierapplications is the base station BTS of FIG. 2. FIG. 6 shows a blockdiagram of possible circuitry for implementing a multi-carrierapplication for amplifying N carriers. The main components of theexemplifying multi-carrier implementation of FIG. 6 correspondsubstantially to the single carrier implementation of FIG. 5 and willthus not be explained in detail. However, combining means 29 arerequired to combine the N modulated signals. The following will discussan exemplifying analysis on which the multi-carrier linearizer may bebased.

A two-carrier input signal, one centred at ω₁ and the other at ω₂, maybe represented in the frequency domain by the following expression:

$\begin{matrix}{{X_{in}\left( {j\;\omega} \right)} = {{{B_{1}\left( {j\;\omega} \right)} \otimes \left\lbrack {{\delta\left( {\omega - \omega_{1}} \right)} + {\delta\left( {\omega + \omega_{1}} \right)}} \right\rbrack} + {{B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {{\delta\left( {\omega - \omega_{2}} \right)} + {\delta\left( {\omega + \omega_{2}} \right)}} \right\rbrack}}} & (8)\end{matrix}$

This signal will produce 6 different in-band third-order intermodulationdistortion components consisting of different combinations of the twobaseband spectrums B₁(jω) and B₂(jω):X _(in)(jω){circle around (×)}X _(in)(jω){circle around (×)}X_(in)(jω)=  (9)

-   -   High-frequency 3^(rd)-order components at 2ω₁+ω₂, 2ω₂+ω₁, 3ω₁,        3ω₂ (these high-frequency 3^(rd)-order components may be easily        removed with filtering, and are thus not written here        explicitly)

$\begin{matrix}\begin{matrix}{{{X_{in}\left( {j\;\omega} \right)} \otimes {X_{in}\left( {j\;\omega} \right)} \otimes {X_{in}\left( {j\;\omega} \right)}} = {{High}\text{-}{frequency}\mspace{14mu} 3^{rd}\text{-}{order}\mspace{14mu}{components}}} \\{{{{at}\mspace{14mu} 2\omega_{1}} + \omega_{2}},\;{{2\omega_{2}} + \omega_{1}},\;{3\omega_{1}},\;{3\omega_{2}}} \\{\left( {{these}\mspace{20mu}{high}\text{-}\left( {{frequency}3} \right)^{rd}\text{-}{order}}\mspace{14mu} \right.} \\{{components}\mspace{14mu}{may}\mspace{14mu}{be}\mspace{14mu}{easily}\mspace{14mu}{removed}} \\{{{{with}\mspace{14mu}{filtering}},{\mspace{11mu}\;}{{and}\mspace{14mu}{are}\mspace{14mu}{thus}\mspace{14mu}{not}}}\mspace{14mu}} \\\left. {{written}\mspace{14mu}{explicitly}} \right) \\{{{+ 3}/8}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{1}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes}} \\{\left\lbrack {{\delta\left( {\omega - \left( {{2\omega_{1}} - \omega_{2}} \right)} \right)} + {\delta\left( {\omega + \left( {{2\omega_{1}} -} \right.} \right.}} \right.} \\{\left. \left. \left. \omega_{2} \right) \right) \right\rbrack + {{3/8}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes}}} \\{{B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {{\delta\left( {\omega - \left( {{2\omega_{2}} - \omega_{1}} \right)} \right)} +} \right.} \\{\delta\left( {\omega + \left( {{2\omega_{2}} -} \right.} \right.} \\{\left. \left. \left. \omega_{1} \right) \right) \right\rbrack + {{3/8}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{1}\left( {j\;\omega} \right)} \otimes}}} \\{\left. \left. {{{B_{1}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega - \omega_{1}} \right)} \right)} + {\delta\left( {\omega + \omega_{1}} \right)}} \right) \right\rbrack +} \\{{3/4}{{B_{1}\left( {j\;\omega} \right)} \otimes}} \\{\left. {{B_{2}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega - \omega_{1}} \right)} \right.} \right) +} \\{\left. \left. {\delta\left( {\omega + \omega_{1}} \right)} \right) \right\rbrack + {{3/8}{{B_{2}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes}}} \\{\left. \left. {{{B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega - \omega_{2}} \right)} \right)} + {\delta\left( {\omega + \omega_{2}} \right)}} \right) \right\rbrack +} \\{{3/4}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{1}\left( {j\;\omega} \right)} \otimes B_{2}}{\left( {j\;\omega} \right) \otimes}} \\\left. \left. {\left\lbrack {\delta\left( {\omega - \omega_{2}} \right)} \right) + {\delta\left( {\omega + \omega_{2}} \right)}} \right) \right\rbrack\end{matrix} & (9)\end{matrix}$

Squaring the input signal in the time domain is equivalent toself-convolution in the frequency domain, therefore:X _(in)(jω){circle around (×)}X _(in)(jω)=  (10)

-   -   High-frequency 2^(nd) order components at (ω₁+ω₂), 2ω₁, 2ω₂ (the        high frequency 2^(nd)-order components are also not shown        explicitly for reasons of brevity)

$\begin{matrix}\begin{matrix}{{{X_{in}({j\omega})} \otimes {X_{in}({j\omega})}} = {{High}\text{-}{frequency}\mspace{14mu} 2^{nd}\text{-}{order}\mspace{14mu}{components}}} \\{{{at}\mspace{14mu}\left( {\omega_{1} + \omega_{2}} \right)},\;{2\omega_{1}},\;{2\omega_{2}\mspace{11mu}\left( {{the}\mspace{20mu}{high}} \right.}} \\{{frequency}\mspace{14mu} 2^{nd}\text{-}{order}\mspace{14mu}{components}\mspace{14mu}{are}{\mspace{14mu}\;}{also}} \\{{not}\mspace{20mu}{shown}\mspace{14mu}{explicity}\mspace{14mu}{for}\mspace{14mu}{reasons}} \\\left. {{of}\mspace{14mu}{brevity}} \right) \\{{{+ 2}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{1}\left( {j\;\omega} \right)} \otimes \delta}(\omega)} +} \\{{2{{B_{2}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \delta}(\omega)} +} \\{2{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega -} \right.} \right.}} \\\left. {\left. \left( {\omega_{2} - \omega_{1}} \right) \right) + {\delta\left( {\omega + \left( {\omega_{2} - \omega_{1}} \right)} \right)}} \right\rbrack\end{matrix} & (10)\end{matrix}$

The last three terms in the above equation (10) represent, in order, theself-convolution of the first baseband, the self-convolution of thesecond baseband (both centred around ω=0) and the convolution of thefirst and second basebands together, centred around ω₂−ω₁ (the frequencyspacing between the two RF carriers assuming ω₂>ω₁). The requiredlinearising signal can thus be generated if the linearising signalgeneration and processing means is supplied with the two basebandwaveforms as well as any appropriate information defining the mutualrelations between the carriers, such as the carrier spacing and/oramplitude information.

Now bulk phase and amplitude adjustment is applied to each of thesethree terms, absorbing the magnitude constants “2” into an amplitudefactor, α.

The generated linearising signal will therefore be:

$\begin{matrix}{{I\left( {j\;\omega} \right)} = {{{\alpha\mathbb{e}}^{- {j\theta}}{B_{1} \otimes B_{1} \otimes {\delta(\omega)}}} + {{\alpha\mathbb{e}}^{- {j\theta}}{{B_{2}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes {\delta(\omega)}}} + {{\alpha\mathbb{e}}^{- {j\theta}}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {{\delta\left( {\omega - \left( {\omega_{2} - \omega_{1}} \right)} \right)} + {\delta\left( {\omega + \left( {\omega_{2} - \omega_{1}} \right)} \right)}} \right\rbrack}}}} & (11)\end{matrix}$

The new input signal (i.e. the 2 original carriers with the newlinearising signal added):

$\begin{matrix}{{{X_{in}}^{\prime}\left( {j\;\omega} \right)} = {{{B_{1}\left( {j\;\omega} \right)} \otimes \left\lbrack {{\delta\left( {\omega - \omega_{1}} \right)} + {\delta\left( {\omega + \omega_{1}} \right)}} \right\rbrack} + {{B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {{\delta\left( {\omega - \omega_{2}} \right)} + {\delta\left( {\omega + \omega_{2}} \right)}} \right\rbrack} + {{\alpha\mathbb{e}}^{- {j\theta}}{B_{1} \otimes B_{1} \otimes {\delta(\omega)}}} + {{\alpha\mathbb{e}}^{- {j\theta}}{{B_{2}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes {\delta(\omega)}}} + {{\alpha\mathbb{e}}^{- {j\theta}}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {{\delta\left( {\omega - \left( {\omega_{2} - \omega_{1}} \right)} \right)} + {\delta\left( {\omega + \left( {\omega_{2} - \omega_{1}} \right)} \right)}} \right\rbrack}}}} & (12)\end{matrix}$

Now we find the self-convolution of this composite input signal (or thesquare of the time-domain equivalent). Again, many components areproduced:

$\begin{matrix}\begin{matrix}{{{{X_{in}}^{\prime}\left( {j\;\omega} \right)} \otimes {{X_{in}}^{\prime}\left( {j\;\omega} \right)}} = \left( {{High}\mspace{11mu}{frequency}\mspace{11mu} 2^{nd}\text{-}{order}\mspace{11mu}{components}\mspace{11mu}{at}}\mspace{11mu} \right.} \\{\left. {{\left( {\omega_{1} + \omega_{2}} \right);\;{2\omega_{1}}},\;{2\omega_{2}}} \right) +} \\{{\alpha\mathbb{e}}^{- {j\theta}}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{1}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega -} \right.} \right.}} \\{\left. {\left. \left( {{2\omega_{1}} - \omega_{2}} \right) \right) + {\delta\left( {\omega + \left( {{2\omega_{1}} - \omega_{2}} \right)} \right)}} \right\rbrack +} \\{{\alpha\mathbb{e}}^{- {j\theta}}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega -} \right.} \right.}} \\{\left. {\left. \left( {{2\omega_{2}} - \omega_{1}} \right) \right) + {\delta\left( {\omega + \left( {{2\omega_{2}} - \omega_{1}} \right)} \right)}} \right\rbrack +} \\{{\alpha\mathbb{e}}^{- {j\theta}}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{1}\left( {j\;\omega} \right)} \otimes {B_{1}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega -} \right.} \right.}} \\{\left. \left. {\left. \left. \omega_{1} \right) \right) + {\delta\left( {\omega + \omega_{1}} \right)}} \right) \right\rbrack +} \\{{\alpha\mathbb{e}}^{- {j\theta}}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega -} \right.} \right.}} \\{\left. \left. {\left. \left. \omega_{1} \right) \right) + {\delta\left( {\omega + \omega_{1}} \right)}} \right) \right\rbrack +} \\{{\alpha\mathbb{e}}^{- {j\theta}}{{B_{2}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega -} \right.} \right.}} \\{\left. \left. {\left. \left. \omega_{2} \right) \right) + {\delta\left( {\omega + \omega_{2}} \right)}} \right) \right\rbrack +} \\{{\alpha\mathbb{e}}^{- {j\theta}}{{B_{1}\left( {j\;\omega} \right)} \otimes {B_{1}\left( {j\;\omega} \right)} \otimes {B_{2}\left( {j\;\omega} \right)} \otimes \left\lbrack {\delta\left( {\omega -} \right.} \right.}} \\\left. \left. {\left. \left. \omega_{2} \right) \right) + {\delta\left( {\omega + \omega_{2}} \right)}} \right) \right\rbrack\end{matrix} & (13)\end{matrix}$

In the same manner as the single-carrier case, it is possible to comparethese new second-order products against those being produced by thethird-order term (9) as shown above. Again, the only difference is theamplitude and phase-shift term, ae^(−j8). This term can be manipulatedby the linearizing signal generation means to cause cancellation of thetwo sets of products. The above can be extended to 3, 4 or any number ofmodulated carriers with a corresponding increase in the complexity ofthe required calculations.

According to a further embodiment also disclosed by FIG. 5 and alreadybriefly discussed above, part or all of the signal that appears at theamplifier output may be analyzed and used to control the function of thelinearizing signal generation circuitry and to monitor the performancethereof. The information obtained from analyzing the output waveform canbe used to maintain the correct linearizing signal amplitude and phaseas operating conditions change and the amplifier ages, should this provenecessary. The linearizing signal 10 may thus be based herein on boththe information from the baseband (signal 8) and the feedbackinformation from the output of the amplifier 4 (signal 11) as well asother information (provided, for example by input 28) as discussedearlier, such as carrier spacing, power levels, temperature and so on.The processing of the signal 11 can be accomplished by the control unit21 or by other appropriate means.

In a preferred form of this further embodiment the feedback control isapplied periodically to adjust the generation of the linearizing signal.By means of this, the arrangement may vary any adjustment parameters ituses to account for any changes in the characteristics and behavior ofthe amplifier. However, the feedback control may also be appliedcontinuously by the control unit 21 to the linearizing signal generationprocess.

Therefore, feedback monitoring and control functionality is notnecessarily dynamic as the generation of the linearizing signal may be,because the controller 21 may act periodically i.e. at intervals. Thefeedback functionality is provided for monitoring the effectiveness ofthe linearizing signal 10 generated by the processing circuit 5 from thebaseband and other relevant information. In case the feedbackfunctionality is not implemented, the arrangement of FIG. 5 may bereferred to as an “open-loop” linearizer. In case the feedbackfunctionality is implemented (e.g. by the feedback connection 11 and thecontroller 21), the implementation may be referred to as a “closed-loop”linearizer.

Digital processing means (filtering or similar) 26 may be used topredistort the injection signal. This predistortion may be necessary insome applications to compensate for the filtering and/or couplingcomponents that must be used in order to combine the separate RF carrierand intermediate frequency (IF) linearizing signals.

According to a more precise embodiment already briefly mentioned in thecontext of FIG. 4, the low-frequency portion of the second-orderdistortion that appears at the amplifier output may be used directlywhen producing the linearizing signal at the signal processing means,such as a data processing device (e.g. a personal computer orworkstation), if the input signal and the low-frequency portion of thesquare of the input signal are periodic waveforms. This may be operatedupon to apply both bulk phase and amplitude adjustment to produce thecorrect linearizing signal as previously outlined, as well as topredistort the linearising signal. The predistortion may be required tocounteract the effects of the low-pass filter networks the linearizingsignal passes through both before being sampled at the output of theamplifier and before being injected back into the input of theamplifier. FIGS. 6 and 7 illustrate as block diagrams the functionalityof two possible methods to apply phase and amplitude correction to theperiodic linearizing signal. FIG. 7 illustrates the use of “Convolution”to process the linearizing signal. FIG. 8 illustrates the use of a“Discrete Fourier-Transform” (DFT) to process the linearizing signal.Briefly, in the convolution method the sampled time waveform ispredistorted by convolving the periodic waveform with animpulse-response representing a transfer function that counteracts theeffects of the low-pass networks discussed, after which the predistortedtime waveform is inverted to adjust the bulk phase of the signal by180°. In the DFT method the captured time waveform is first translatedinto the frequency domain by being subjected to a Discrete FourierTransform at specifiable “spot” frequencies which are equal to theselected carrier spacings (see s1–s6 on FIG. 4). The results of thistime-to-frequency transform are the relative amplitude and phase valuesof each signal component present. A previously collected look-up tableof frequency-response data is then referenced to adjust the phase andamplitude of each component, thus counteracting the effect of thelow-pass networks already discussed. Unlike the convolution method thatuses a best-fit impulse-response to predistort the injection signal, theDFT allows very precise adjustment of the amplitude and phase of eachindividual frequency component within the linearising signal. It isnoted that a Fast Fourier Transform (FFT) may be employed in place of aDiscrete Fourier Transform to effect frequency-domain analysis of bothperiodic and non-periodic signals.

Embodiments of the invention have also been subjected to experimentaltests and simulations. Results of some of the tests performed forfour-tone continuous waves (CW) test signals are shown by the diagramsof FIGS. 9 to 11. The carrier spacings were selected as 1 MHz, 4 MHz and3 MHz, respectively, giving 6 second-order difference frequencycomponents 1 MHz, 3 MHz, 4 MHz, 5 MHz, 7 MHz and 8 MHz (for the 6different carrier spacings, see FIG. 4). The multi-tone test bench wasprovided with a computer control unit utilizing utilizing VisualBasic toenable data acquisition, waveform processing and basic DSPfunctionality. The four test-tones were generated by respective signalgenerators, then pre-amplified and combined in a separate pre-amplifyingand combining unit prior to being inputted into a power amplifier (thedevice under test was a Fujitsu FLL351 ME FET). The injection of thelinearizing signal was implemented by means of an arbitrary waveformgenerator via an inductive bias-feeding line. The IF waveform consistingof the low-frequency portion of the square of the RF input signal wascaptured at the output of the amplifier with a sampling oscilloscopealso via an inductive DC biasfeed line. The amplified output signal wasanalyzed and recorded by a spectrum analyzer.

These tests demonstrated that injection of a linearizing IF signal canreduce distortion in the RF band with any number of continuous wave (CW)input signals. FIG. 9 shows the result of a 4-tone input signal withequal-power carriers and with the above-discussed DFT predistortionemployed in the generation of the linearizing signal. The resultobtained when the low-frequency linearizing signal was not added to theRF input signal is shown by a dashed line and the result obtained withthe linearizing signal added is shown by a solid line. FIGS. 9 and 10illustrate results for similar tests, but for carriers with differentpowers, FIG. 11 illustrating a situation where the power of the tones is3 dB greater than those shown in FIG. 10.

The embodiment of FIG. 5 was subjected to simulation, FIG. 12 being aschematic representation of the circuitry constructed for thesimulations. It is noted that while the CW 4-tone results discussedabove were obtained through practical experiments, the circuit of FIG.12 was built and tested with a computer simulation package (“AdvancedDesign System™” from Hewlett Packard). The simulation results obtainedfor a modulated carrier are shown in FIGS. 13 and 14. FIG. 13 shows thefrequency spectrum of the output of the amplifier when the linearizingsignal was not added to the input signal, while FIG. 14 shows thespectrum of the output of the amplifier with the same input carriersignal, modified by the addition of the required linearizing signal.

With reference to FIG. 12, the simulations were accomplished usingdigital baseband signals consisting of two separate signals,commonly-known as “I” and “Q” for “in-phase” and “quadrature-phase”,respectively. These were 2 separate streams of digital data thatrepresent the baseband information when they are used together. Thebaseband information was separated in this known manner because itenables twice as much information to be carried in the same bandwidth.The digital streams were low-pass filtered to restrict their bandwidth,also in a known manner. Mixers 12—performed the ‘upconversion’ from IFto RF by each multiplying one of the filtered baseband waveforms by ahigh-frequency sinusoidal carrier, translating them up to 800 MHz. Asinusoidal 800 MHz local-oscillator (LO) divided the signal into twobranches. The branch that was to become the quadrature half of themodulated carrier was subjected to 90 degrees of phase-shift. The mixerswere also provided with upconverted RF output connections. The I and Qhalves of the signal were then combined in 31 to form the complete RFcarrier. The carrier was filtered to remove extraneous signals producedby the upconversion process by a bandpass filter shown connected betweenunits 31 and 15.

The two filtered bit-streams were also provided to the processingcircuitry 5 via connections 81 and 8Q, respectively, and the linearizingsignal was generated from these bit-streams in the following manner. Thebaseband I and Q signals were squared before combination. In order toimplement this in the simulator, the signal was split and the two halvesmultiplied together (this could be realized in the same manner in therespective signal processors 5 of FIG. 4 or 5). Following the squaringfunction the simulation arrangement comprised means for adjusting thebulk amplitude and phase of the linearizing signal, labelled 24 and 25respectively. The IF injection signal 10 was then added to theupconverted carrier using the idealized combining means represented bythe block 15. Finally, the signal was inputted into the realisticnonlinear amplifier model 34 that was to be linearizing.

When comparing the results of FIGS. 13 and 14, it can be seen that adistortion reduction can be achieved by the injection of the requiredlinearizing a signal. The adjacent channel power ratio (ACPR) around thecarrier is clearly at a lower level in FIG. 14 than in FIG. 13.

In the accomplished tests and simulations the distortion reductionvaried within a range of 10 dB to about 30 dB. However, even biggerimprovements in the cancellation of the intermodulation distortion arebelieved to be possible by the embodiments of the invention.

Even though not tested yet, it is believed that an improvement is alsoobtainable by the embodiments in cancellation of AM-PM distortion.

It should be appreciated that the above embodiments can also be used inconjunction with other types of linearizers to relax manufacturingtolerances and/or improve overall performance.

It should be appreciated that whilst embodiments of the presentinvention have been described in relation to stations of a radiocommunication system, embodiments of the present invention areapplicable to any other suitable type of transmitter and/or receiverequipment.

It is also noted herein that while the above describes exemplifyingembodiments of the invention, there are several variations andmodifications which may be made to the disclosed solution withoutdeparting from the scope of the present invention as defined in theappended claims.

1. A method of linearizing an amplifier, the amplifier being providedfor amplifying a signal, the method comprising steps of: generating alinearizing signal that substantially corresponds to low-frequencycomponents of square of an input signal of the amplifier; processing thelinearizing signal; combining the linearizing signal with the inputsignal to form a modified input signal; and amplifying the modifiedinput signal by means of the amplifier, wherein the step of processingthe linearizing signal comprises applying amplitude adjustment andphase-shift to the generated linearizing signal, wherein the amplitudeof the linearizing signal is chosen such that the amplitude ofthird-order distortion components generated by a second-order distortionof the amplifier and the amplitude of pre-existing third-orderdistortion components around a carrier signal are equal.
 2. A methodaccording to claim 1, wherein the phase-shift applied to the linearizingsignal is such that the phase of the third-order distortion componentsgenerated by the second-order distortion of the amplifier is opposite tothe phase of the pre-existing third-order distortion around the carriersignal.
 3. A method according to claim 2, wherein intermodulationdistortion is reduced at the output of the amplifier by means ofaddition of the linearizing signal to the input signal.
 4. A methodaccording to claim 3, wherein the intermodulation distortion comprisesthird-order intermodulation distortion in a radio frequency poweramplifier.
 5. A method according to claim 3, wherein the intermodulationdistortion comprises fifth-order intermodulation distortion.
 6. A methodaccording to claim 5, wherein the linearizing signal is combined withthe input signal after the input signal is converted from intermediatefrequency to radio frequency.
 7. A method of linearizing an amplifier,the amplifier being provided for amplifying a signal, the methodcomprising steps of: generating a linearizing signal that substantiallycorresponds to the low-frequency components of the square of an inputsignal to the amplifier; processing the linearizing signal; combiningthe linearizing signal with the input signal to form a modified inputsignal; and amplifying the modified input signal by means of amplifier,wherein the input signal consists of two or more modulated carriersignals comprising squaring two or more baseband waveforms and usinginformation associated with the mutual relations between said carriersignal, wherein the information concerning the mutual relationscomprises carrier spacings between said two or more carrier signal. 8.Circuitry comprising: an amplifier provided with an input for receivinga signal; processing means for generating a linearizing signal thatsubstantially corresponds to low-frequency components of square of aninput signal for the amplifier and for processing the linearizingsignal; and combiner means for combining the linearizing signal with theinput signal to form a modified input signal to be applied to the inputof the amplifier for amplification, wherein the processing meanscomprises a digital signal processing circuit, wherein the signalprocessing circuit is arranged to square baseband waveform of the inputsignal and to apply amplitude and phase-shift to the low-frequencycomponents resulting from said squaring.
 9. Circuitry according to claim8, wherein the amplifier comprises a radio frequency power amplifier andthe intermodulation distortion consisting of third-order and/orfifth-order intermodulation distortion around the input signal.